EC1837|80V/2A Step-Down Converter


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EC1837

80V/2A Step-Down Converter

General Description

The EC1837 is a high frequency step-down switching regulator with integrated internal highside high voltage power MOSFET. It provides 2A output with current mode control for fast loop response and easy compensation. The wide 12V to 80V input range accommodates a variety of step-down applications, including those in automotive input environment. A 1μA shutdown mode supply current allows use in battery-powered applications. High power conversion efficiency over a wide load range is achieved by scaling down the switching frequency at light

load condition to reduce the switching and gate driving losses. The frequency foldback helps prevent inductor current runaway during startup and thermal shutdown provides reliable, fault tolerant operation.

The EC1837 is available in ESOP8 package.

 

Features

Wide 12V to 80V Operating Input Range

250mΩ Internal Power MOSFET

Up to 1MHz Programmable Switching Frequency

180μA Quiescent Current

Ceramic Capacitor Stable

Internal Soft-Start

Up to 95% Efficiency

Output Adjustable from 0.8V to 52V

Available in ESOP8 PbFree Package

 

Applications

High Voltage Power Conversion

Automotive Systems

Industrial Power Systems

Distributed Power Systems

Battery Powered Systems

 

Pin Configurations

(Top view)

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EC1837

 

 

80V/2A Step-Down Converter

Pin Description

Pin Number

Pin Name

Description

 

1

 

SW

Switch Node. This is the output from the high-side switch. A low VF Schottky rectifier to ground

is required. The rectifier must be close to the SW pins to reduce switching spikes.

 

2

 

EN

Enable Input. Pulling this pin below the specified threshold or leaving it floating shuts the chip

down. Pulling it up above the specified threshold enables the chip.

 

3

 

COMP

Compensation. This node is the output of the GM error amplifier. Control loop frequency

compensation is applied to this pin.

 

4

 

FB

Feedback. This is the input to the error amplifier. An external resistive divider connected between the output and GND is compared to the internal +0.8V reference to set the regulation voltage.

 

5

GND,

Exposed

pad

 

Ground. It should be connected as close as possible to the output capacitor avoiding the high current switch paths. Connect exposed pad to GND plane for optimal thermal performance.

 

6

 

FREQ

Switching Frequency Program Input. Connect a resistor from this pin to ground to set the

switching frequency.

 

7

 

VIN

Input Supply. This supplies power to all the internal control circuitry, both BS regulators and the high-side switch. A decoupling capacitor to ground must be placed close to this pin to minimize switching spikes.

 

8

 

BST

Bootstrap. This is the positive power supply for the internal floating high-side MOSFET driver.

Connect a bypass capacitor between this pin and SW pin.

 

 

Ordering Information

 

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EC1837NN ​​ XX ​​ X

 

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Part Number

Package

Marking

Marking Information

EC1837NNM1R

ESOT8

EC1837

                LLLLL

                YYWW

LLLLL is Lot Number

YYWW is date code

 

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EC1837

 

 

80V/2A Step-Down Converter

Function Block

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

Figure1 Function Block Diagram of EC1837

 

Typical Application Circuit

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

 

Figure2 Application Circuit, 5V Output

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EC1837

 

 

80V/2A Step-Down Converter

Absolute Maximum Ratings (at TA=25℃)

Symbol

Parameter

Rating

Unit

VIN

Supply Voltage

-0.3 to 80

V

VSW

Switch Voltage

-0.5V to VIN+0.5

V

 

BST to SW

-0.3 to +5

V

 

All other Pins

-0.3 to +5

V

 

ESD

 

ESD Susceptibility (Human Body Model)

 

2K

V

TJ

Junction Temperature

150

ºC

 

TSDR

 

Maximum Lead Soldering Temperature (10 Seconds)

 

260

ºC

 

Electrical Characteristics

Unless otherwise specified, these specifications apply over VIN=24V,VEN=2.5V,VCOMP=1.4V, TA=25°C

Specifications over temperature are guaranteed by design and characterization.

Characteristics

Symbol

Conditions

Min

Typ

Max

Units

Feedback Voltage

VFB

12V < VIN < 75V

0.780

0.800

0.820

V

Top Switch RDS(ON) (Note)

RDS(ON)-T

VBST VSW = 5V

175

250

330

mΩ

Top Switch Leakage

 

VEN = 0V, VSW = 0V

-

1

-

μA

Current Limit

 

 

2.2

-

4.7

A

COMP to Current Sense

Transconductance

 

GCS

 

-

5.7

-

A/V

Error Amp Voltage Gain

 

 

-

400

-

V/V

Error Amp Transconductance

 

ICOMP = ±3μA

-

120

-

μA/A

Error Amp Min Source current

 

VFB = 0.7V

-

10

-

μA

Error Amp Min Sink current

 

VFB = 0.9V

-

-10

-

μA

VIN UVLO Threshold

 

 

-

7.2

-

V

VIN UVLO Hysteresis

 

 

-

0.5

-

V

Soft-Start Time

 

0V < VFB < 0.8V

-

0.5

-

ms

Oscillator Frequency

 

RFREQ = 150kΩ

0.6

-

0.8

MHz

Minimum Switch On Time

 

 

-

100

-

ns

Shutdown Supply Current

 

VEN < 0.3V

-

1

3

μA

Quiescent Supply Current

 

No load, VFB = 0.9V

-

180

-

μA

Thermal Shutdown

 

Hysteresis = 20°C

-

150

-

°C

Minimum Off Time

 

 

-

100

-

ns

Minimum On Time

 

 

-

100

-

ns

EN Up Threshold

 

 

1.3

-

2.2

V

EN Threshold Hysteresis

 

 

-

200

-

mV

EN to Gnd resistance

 

 

-

1

-

MΩ

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EC1837

 

 

80V/2A Step-Down Converter

Typical Efficiency

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EC1837

 

 

80V/2A Step-Down Converter

Operation

The EC1837 is a programmable frequency, non-synchronous, step-down switching regulator with an integrated high-side high ​​ voltage ​​ power ​​ MOSFET. ​​ It ​​ provides ​​ a ​​ single ​​ highly efficient solution with current mode control for fast loop response and easy compensation. It features a wide input voltage ​​ range, ​​ internal ​​ soft-start ​​ control ​​ and ​​ precision current limiting. Its very low operational quiescent current makes it suitable for battery powered applications.

 

PWM Control Mode

At moderate to high output current, the EC1837 operates in a fixed frequency, peak current control mode to regulate the output voltage. A PWM cycle is initiated by the internal clock. The power MOSFET is turned on and remains on until its ​​ current ​​ reaches ​​ the ​​ value ​​ set ​​ by ​​ the ​​ COMP ​​ voltage. When the power switch is off, it remains off for at least 100ns before the next cycle starts. If, in one PWM period, the current in the power MOSFET does not reach the COMP set current value, the power MOSFET remains on, saving a turn-off operation.

 

Pulse Skipping Mode

Under light load condition the switching frequency stretches down zero to reduce the switching loss and driving loss.

 

Error Amplifier

The error amplifier compares the FB pin voltage with the internal reference (REF) and outputs a current proportional to the difference between the two. This output current is then used to charge the external compensation network to form the COMP voltage, which is used to control the power MOSFET current.

During operation, the minimum COMP voltage is clamped to 0.9V ​​ and ​​ its ​​ maximum ​​ is ​​ clamped ​​ to ​​ 2.0V. ​​ COMP ​​ is internally pulled down to GND in shutdown mode. COMP should not be pulled up beyond 2.6V.

 

Internal Regulator

Most of the internal circuitries are powered from the 2.6V internal regulator. This regulator takes the VIN input and operates in the full VIN range. When VIN is greater than 3.0V, the output of the regulator is in full regulation. When VIN is lower than 3.0V, the output decreases.

 

Enable Control

The EC1837 has a dedicated enable control pin(EN). With high ​​ enough input voltage, ​​ the ​​ chip ​​ can be ​​ enabled and disabled by EN which has positive logic. Its falling threshold is about 1.7V, and its rising threshold is about 1.9V.

When EN is pulled down below 1.7V, the chip is put into the lowest shutdown ​​ current mode. ​​ When EN ​​ is higher than zero but lower than its rising threshold, the chip is still in shutdown  ​​​​ mode  ​​​​ but  ​​​​ the  ​​​​ shutdown  ​​​​ current  ​​​​ increases slightly.

 

Under-Voltage Lockout (UVLO)

Under-voltage lockout (UVLO) is implemented to protect the chip from operating at insufficient supply voltage. The UVLO rising threshold is about 7.2V while its falling threshold is a consistent 6.5V.

 

Internal Soft-Start

The ​​ soft-start ​​ is ​​ implemented ​​ to ​​ prevent ​​ the ​​ converter output voltage from overshooting during startup and short circuit recovery. When the chip starts, the internal circuitry generates a soft-start voltage (SS) ramping up from 0V to 2.6V. When it is lower than the internal reference(REF), SS overrides REF so the error amplifier uses SS as the reference. When SS is higher than REF, REF regains control.

 

Thermal Shutdown

Thermal shutdown is implemented to prevent the chip from operating ​​ at ​​ exceedingly ​​ high ​​ temperatures. ​​ When ​​ the silicon die temperature is higher than its upper threshold, it shuts down the whole chip. When the temperature is lower than its lower threshold, the chip is enabled again.

 

Floating Driver and Bootstrap Charging

The ​​ floating ​​ power ​​ MOSFET ​​ driver ​​ is ​​ powered ​​ by ​​ an external bootstrap capacitor. This floating driver has its own UVLO protection. This UVLO’s rising threshold is 7.2V with a hysteresis of 150mV. The driver’s UVLO is soft-start related.

In case the bootstrap voltage hits its UVLO, the soft-start circuit is reset. To prevent noise, there is 20μs delay before the reset action. When bootstrap UVLO is gone, the reset is off and then soft-start process resumes.

The bootstrap capacitor is charged and regulated to about 5V by the dedicated internal bootstrap regulator. When the voltage between the BST and SW nodes is lower than its regulation, a PMOS pass transistor connected from VIN to BST is turned on. The charging current path is from VIN, BST and ​​ then ​​ to ​​ SW. ​​ External ​​ circuit ​​ should ​​ provide ​​ enough voltage headroom to facilitate the charging.

As long as VIN is sufficiently higher than SW, the bootstrap capacitor can be charged. When the power MOSFET is ON, VIN is about equal to SW so the bootstrap capacitor cannot be charged. When the external diode is on, the difference between ​​ VIN ​​ and ​​ SW ​​ is ​​ largest, ​​ thus ​​ making ​​ it ​​ the ​​ best period to charge. When there is no current in the inductor, SW ​​ equals ​​ the ​​ output ​​ voltage ​​ VOUT ​​ so ​​ the ​​ difference between VIN and VOUT can be used to charge the bootstrap capacitor.

At higher duty cycle operation condition, the time period available to the bootstrap charging is less so the bootstrap capacitor may not be sufficiently charged.

In case the internal circuit does not have sufficient voltage and the bootstrap capacitor is not charged, extra external circuitry can be used to ensure the bootstrap voltage is in the normal operational region. Refer to External Bootstrap Diode in Application section.

The ​​ DC ​​ quiescent ​​ current ​​ of ​​ the ​​ floating ​​ driver ​​ is ​​ about 20μA. Make sure the bleeding current at the SW node is higher than this value, such that:

 

 

 

Current Comparator and Current Limit

The ​​ power ​​ MOSFET ​​ current ​​ is ​​ accurately ​​ sensed ​​ via ​​ a current ​​ sense ​​ MOSFET. ​​ It ​​ is ​​ then ​​ fed ​​ to ​​ the ​​ high ​​ speed current comparator for the current mode control purpose. The current comparator takes this sensed current as one of its ​​ inputs. ​​ When ​​ the ​​ power ​​ MOSFET ​​ is ​​ turned ​​ on, ​​ the comparator ​​ is ​​ first ​​ blanked ​​ till ​​ the ​​ end ​​ of ​​ the ​​ turn ​​ on transition ​​ to ​​ avoid ​​ noise ​​ issues. ​​ The ​​ comparator ​​ then compares the power switch current with the COMP voltage.

When the sensed current is higher than the COMP voltage, the ​​ comparator ​​ output ​​ is ​​ low, ​​ turning ​​ off ​​ the ​​ power MOSFET.  ​​​​ The  ​​​​ cycle-by-cycle  ​​​​ maximum  ​​​​ current  ​​​​ of  ​​​​ the internal power MOSFET is internally limited.

 

Short Circuit Protection

When the output is shorted to the ground, the switching frequency is folded back and the current limit is reduced to lower the short circuit current. When the voltage of FB is at zero, the current limit is reduced to about 50% of its full current limit. When FB voltage is higher than 0.4V, current limit reaches 100%.

In short circuit FB voltage is low, the SS is pulled down by FB and SS is about 100mV above FB. In case the short circuit is removed, the output voltage will recover at the SS pace. When FB is high enough, the frequency and current limit return to normal values.

 

Startup and Shutdown

If ​​ both ​​ VIN ​​ and ​​ EN ​​ are ​​ higher ​​ than ​​ their ​​ appropriate thresholds, the chip starts. The reference block starts first, generating stable reference voltage and currents, and then the ​​ internal ​​ regulator ​​ is ​​ enabled. ​​ The ​​ regulator ​​ provides stable supply for the remaining circuitries.

While the internal supply rail is up, an internal timer holds the power MOSFET OFF for about 50μs to blank the startup glitches. When the internal soft-start block is enabled, it first holds its SS output low to ensure the remaining circuitries are ready and then slowly ramps up.

Three events can shut down the chip: EN low, VIN low and thermal ​​ shutdown. ​​ In ​​ the ​​ shutdown ​​ procedure, ​​ power MOSFET is turned off first to avoid any fault triggering. The COMP voltage and the internal supply rail are then pulled down.

 

Programmable Oscillator

The ​​ EC1837 oscillating ​​ frequency ​​ is ​​ set ​​ by ​​ an ​​ external resistor, RFREQ from the FREQ pin to ground. The value of RFREQ can be calculated from:

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EC1837

 

 

80V/2A Step-Down Converter

Application information

Setting the Output Voltage

The output voltage is set using a resistive voltage divider from ​​ the ​​ output ​​ voltage ​​ to ​​ FB ​​ pin. ​​ The ​​ voltage ​​ divider divides the output voltage down to the feedback voltage by the ratio:

Thus the output voltage is:

 

For example, the value for R2 can be 10kΩ. With this value, R1 can be determined by:

For example, for a 3.3V output voltage, R2 is 10kΩ, and R1 is 31.6kΩ.

 

Inductor

The inductor is required to supply constant current to the output ​​ load ​​ while ​​ being ​​ driven ​​ by ​​ the ​​ switched ​​ input voltage. ​​ A ​​ larger ​​ value ​​ inductor ​​ will ​​ result ​​ in ​​ less ​​ ripple current ​​ that ​​ will ​​ result ​​ in ​​ lower ​​ output ​​ ripple ​​ voltage. However, ​​ the ​​ larger ​​ value ​​ inductor ​​ will ​​ have ​​ a ​​ larger physical size, higher series resistance, and/or lower saturation current.

A ​​ good ​​ rule ​​ for ​​ determining ​​ the ​​ inductance ​​ to ​​ use ​​ is ​​ to allow the peak-to-peak ripple current in the inductor to be approximately 30% of the maximum switch current limit. Also, make sure that the peak inductor current is below the maximum switch current limit. The inductance value can be calculated by:

 

Where VOUT is the output voltage, VIN is the input voltage, fS is the switching frequency, and ΔIL is the peak-to-peak inductor ripple current.

Choose  ​​​​ an  ​​​​ inductor  ​​​​ that ​​ will  ​​​​ not  ​​​​ saturate  ​​​​ under  ​​​​ the maximum inductor peak current. The peak inductor current can be calculated by:

 

 

Where ILOAD is the load current.

Table 1 lists a number of suitable inductors from various manufacturers. The choice of which style inductor to use mainly depends on the price vs. size requirements and any EMI requirement.

 

Output Rectifier Diode

The ​​ output ​​ rectifier ​​ diode ​​ supplies ​​ the ​​ current ​​ to ​​ the inductor when the high-side switch is off. To reduce losses due to the diode forward voltage and recovery times, use a Schottky diode.

Choose a diode whose maximum reverse voltage rating is greater  ​​​​ than  ​​​​ the  ​​​​ maximum  ​​​​ input ​​ voltage,  ​​​​ and ​​ whose current rating is greater than the maximum load current. Table 2 lists example Schottky diodes and manufacturers.

 

 

 

 

 

 

 

 

Input Capacitor

The input current to the step-down converter is discontinuous, therefore a capacitor is required to supply the AC current to the step-down  converter while maintaining the DC input voltage. Use low ESR capacitors for the best performance. Ceramic capacitors are preferred, but tantalum or low-ESR electrolytic capacitors may also suffice. For ​​ simplification, choose the input capacitor with RMS current rating greater than half of the maximum load current.The input capacitor (C1) can be electrolytic, tantalum or ceramic.

When using electrolytic or tantalum capacitors, a small, high quality ceramic capacitor, i.e. 0.1μF, should be placed as close to the IC as possible. When using ceramic capacitors,

make sure that they have enough capacitance to provide sufficient charge to prevent excessive voltage ripple at input. The ​​ input ​​ voltage ​​ ripple ​​ caused ​​ by ​​ capacitance ​​ can ​​ be estimated by:

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EC1837

 

 

80V/2A Step-Down Converter

Output Capacitor

The output capacitor (C2) is required to maintain the DC output voltage. Ceramic, tantalum, or low ESR electrolytic capacitors  ​​​​ are  ​​​​ recommended.  ​​​​ Low  ​​​​ ESR  ​​​​ capacitors  ​​​​ are preferred to keep the output voltage ripple low. The output voltage ripple can be estimated by:

 

 

Where L is the inductor value and RESR is the equivalent series resistance (ESR) value of the output capacitor.

In ​​ the ​​ case ​​ of ​​ ceramic ​​ capacitors, ​​ the ​​ impedance ​​ at ​​ the switching frequency is dominated by the capacitance. The output voltage ripple is mainly caused by the capacitance. For  ​​​​ simplification,  ​​​​ the  ​​​​ output  ​​​​ voltage  ​​​​ ripple  ​​​​ can  ​​​​ be estimated by:

 

 

In the case of tantalum or electrolytic capacitors, the ESR dominates the impedance at the switching frequency. For simplification, the output ripple can be approximated to:

 

 

The characteristics of the output capacitor also affect the stability ​​ of ​​ the ​​ regulation ​​ system. ​​ The EC1837 can be optimized for a wide range of capacitance and ESR values.

 

Compensation Components

EC1837 employs current mode control for easy compensation ​​ and ​​ fast ​​ transient ​​ response. ​​ The ​​ system stability and transient response are controlled through the COMP pin. COMP pin is the ​​ output of the ​​ internal error amplifier. ​​ A ​​ series ​​ capacitor-resistor ​​ combination ​​ sets ​​ a pole-zero combination to control the characteristics of the control system. The DC gain of the voltage feedback loop is given by:

 

 

Where AVEA is the error amplifier voltage gain, 400V/V; GCS is the current sense transconductance, 5.6A/V; RLOAD is the load resistor value.

The system has two poles of importance. One is due to the compensation capacitor (C3), the output resistor of error amplifier. The other is due to the output capacitor and the load resistor. These poles are located at:

 

 

 

 

Where,  ​​​​ GEA  ​​​​ is  ​​​​ the  ​​​​ error  ​​​​ amplifier  ​​​​ transconductance, 120μA/V.

The ​​ system ​​ has ​​ one ​​ zero ​​ of ​​ importance, ​​ due ​​ to ​​ the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at:

 

 

 

The system may have another zero of importance, if the output capacitor has a large capacitance and/or a high ESR value. ​​ The ​​ zero, ​​ due ​​ to ​​ the ​​ ESR ​​ and ​​ capacitance ​​ of ​​ the output capacitor, is located at:

 

 

In this case, a third pole set by the compensation capacitor (C5)  ​​​​ and  ​​​​ the  ​​​​ compensation  ​​​​ resistor  ​​​​ (R3)  ​​​​ is  ​​​​ used  ​​​​ to compensate the effect of the ESR zero on the loop gain. This pole is located at:

 

 

The goal of compensation design is to shape the converter transfer ​​ function ​​ to ​​ get ​​ a ​​ desired ​​ loop ​​ gain. ​​ The ​​ system crossover frequency where the feedback loop has the unity gain ​​ is ​​ important. ​​ Lower ​​ crossover ​​ frequencies ​​ result ​​ in slower ​​ line ​​ and ​​ load ​​ transient ​​ responses, ​​ while ​​ higher crossover frequencies could cause system unstable. A good rule  ​​​​ of  ​​​​ thumb  ​​​​ is  ​​​​ to  ​​​​ set  ​​​​ the  ​​​​ crossover  ​​​​ frequency  ​​​​ to approximately one-tenth of the switching frequency.

To optimize the compensation components for conditions not listed in Table 3, the following procedure can be used.

1. Choose the compensation resistor (R3) to set the desired crossover  ​​​​ frequency.  ​​​​ Determine  ​​​​ the  ​​​​ R3  ​​​​ value  ​​​​ by  ​​​​ the following equation:

 

Where fC is the desired crossover frequency.